1. Field of the Invention
The present invention relates to a DC-DC converter, and particularly, to feedback control of the DC-DC converter.
2. Description of the Related Art
FIG. 1 is a circuit diagram illustrating a DC-DC converter according to a related art. Operation of the DC-DC converter of FIG. 1 will be explained. A DC power source Vin applies a voltage to a starter (not illustrated) to start a controller 10. The controller 10 has an oscillator 11, a D-flip-flop 13, dead time generators 14 and 15, a level shifter 16, and buffers 17 and 18, to alternately turn on/off switch elements Q1 and Q2 with a dead time included.
When the switch element Q2 is turned on, a current passes clockwise through a path extending along Vin, Q2, Lr, P, Cri, and Vin. This current is a resultant current of an excitation current passing through an excitation inductance Lp on the primary side of a transformer T and a load current supplied through a primary winding P, secondary winding S2, diode D2, and a capacitor Co to output terminals +Vo and −Vo to a load. The excitation current is a sinusoidal resonant current created by an inductive reactance of the reactor Lr and the excitation inductance Lp and a capacitive reactance of the current resonant capacitor Cri. To make the frequency of the sinusoidal resonant current lower than an ON period of the switch element Q2, the sinusoidal wave of the resonant current is partly observed as a triangular wave. The load current is a sinusoidal resonant current created by resonant components of the reactor Lr and current resonant capacitor Cri.
When the switch element Q2 is turned off, energy accumulated in the transformer T by the excitation current causes a quasi-voltage-resonance of the inductive reactance to the reactor Lr and the excitation inductance Lp and the capacitive reactance to the current resonant capacitor Cri and a voltage resonant capacitor Cry. At this time, a resonant frequency by the voltage resonant capacitor Cry of small capacitance is observed as a voltage across the switch elements Q1 and Q2. Namely, a current of the switch element Q2 is switched when the switch element Q2 is turned off to a current passing through the voltage resonant capacitor Crv. When the voltage resonant capacitor Cry is discharged to 0 V level, the current path is switched to a diode D3. Then, the energy accumulated in the transformer T by the excitation current charges the current resonant capacitor Cri through the diode D3. During this period, the switch element Q1 is turned on to realize zero voltage switching of the switch element Q1.
When the switch element Q1 is turned on, the current resonant capacitor Cri serves as a power source to pass a current counterclockwise through a path extending along Cri, P, Lr, Q1, and Cri. This current is a resultant current of an excitation current passing through the excitation inductance Lp of the transformer T and a load current supplied through the primary winding P, secondary winding S1, diode D1, and smoothing capacitor Co to the output terminals +Vo and −Vo to the load. The excitation current is a sinusoidal resonant current created by the reactor Lr, the excitation inductance Lp, and the current resonant capacitor Cri. To make the frequency of the sinusoidal resonant current lower than an ON period of the switch element Q1, the sinusoidal wave of the resonant current is partly observed as a triangular wave. The load current is a sinusoidal resonant current created by resonant components of the reactor Lr and current resonant capacitor Cri.
When the switch element Q1 is turned off, energy accumulated in the transformer T by the excitation current causes a quasi-voltage-resonance of the inductive reactance to the reactor Lr and the excitation inductance Lp and the capacitive reactance to the current resonant capacitor Cri and the voltage resonant capacitor Cry. At this time, a resonant frequency by the voltage resonant capacitor Cry of small capacitance is observed as a voltage across the switch elements Q1 and Q2. Namely, a current of the switch element Q1 is switched when the switch element Q1 is turned off to a current passing through the voltage resonant capacitor Cry. When the voltage resonant capacitor Cry is charged to the voltage of the DC power source Vin, the current path is switched to a diode D4. This means that the energy accumulated in the transformer T by the excitation current is regenerated through the diode D4 to the DC power source Vin. During this period, the switch element Q2 is turned on to realize zero voltage switching of the switch element Q2.
FIG. 2A illustrates waveforms at characteristic parts of the DC-DC converter of FIG. 1 with the DC power source Vin being 300 V, the load being 100% (heavy load), and the switching frequency being 43.1 kHz and FIG. 2B illustrates waveforms at the essential parts with the DC power source Vin being 450 V, the load being 100%, and the switching frequency being 74.6 kHz. By comparing FIGS. 2A and 2B with each other, one can grasp changes that occur at the essential parts when the input voltage varies under heavy load.
In this example, the controller 10 fixes the dead time, and according to the varying input voltage, controls the switching frequency to alternately turn on/off the switch elements Q1 and Q2. On an assumption that the frequency of the resonant current to the load is constant, the controller 10 controls the switching frequency to, for example, widen an ON width and increase the excitation current that is a circulation current, thereby changing the voltage amplitude of the current resonant capacitor Cri and controlling an output voltage.
FIG. 2C illustrates waveforms at the essential parts with the DC power source Vin being 300 V, the load being 0.01% (no load), and the switching frequency being 47.1 kHz and FIG. 2D illustrates waveforms at the essential parts with the DC power source Vin being 450 V, the load being 0.01% (no load), and the switching frequency being 83.3 kHz. By comparing FIGS. 2C and 2D with each other, one can grasp changes that occur at the essential parts when the input voltage varies under no load.
By comparing FIGS. 2A and 2C, or FIGS. 2B and 2D with each other, one can grasp changes that occur at the essential parts when the load varies under the same input voltage. The waveforms of FIG. 2A involve a resonant current corresponding to a load current because of the heavy-load condition. The waveforms of FIG. 2C involve substantially no resonant current corresponding to a load current because of the no-load condition. It is understood from the waveforms of FIGS. 2A and 2C that the switching frequency is substantially unchanged with respect to a load variation.
An actual DC-DC converter involves many inductances and capacitances that are not illustrated in FIG. 1. Generally, they cause no large influence on operation of the DC-DC converter. There is, however, some instance in which the inductances and capacitances are not ignorable. For example, the transformer T of FIG. 1 has an inter-winding stray capacitance Cm indicated with a dotted line in FIG. 1. If the inter-winding stray capacitance Cm is relatively large, it affects operation of the DC-DC converter.
FIGS. 3A to 3D, which correspond to FIGS. 2A to 2D, illustrate waveforms at the characteristic parts of the DC-DC converter including the inter-winding stray capacitance. Unlike the waveforms of FIGS. 2A to 2D, the waveforms of FIGS. 3A to 3D show a large change in the switching frequency when the load varies. This is because the inter-winding stray capacitance Cm and winding inductance create an oscillating voltage and because the peak voltage thereof is peak-charged at the time of rectification.
The peak charging tends to increase a voltage after rectification, and therefore, a feedback circuit (a photocoupler PC) provides a large feedback amount to the controller 10. Due to this, the controller 10 increases the oscillation frequency (corresponding to the switching frequency) of the oscillator 11, to suppress an increase in the output voltage V0. Namely, the feedback amount is small in the range from heavy to light load and is large in the range from light to no load. FIG. 4 illustrates a relationship between an output power ratio and a switching frequency. As is apparent in FIG. 4, the presence of the inter-winding stray capacitance increases the switching frequency as the output power ratio decreases.
FIG. 5 illustrates a relationship between the feedback current and oscillation frequency of the controller 10 of the DC-DC converter according to the related art. As is apparent in FIG. 5, the feedback current and oscillation frequency have a proportional relationship. When the inter-winding stray capacitance is small, there is a little change in the oscillation frequency, so that a small feedback current is sufficient to control the DC-DC converter.
If the inter-winding stray capacitance is large, a change in the oscillation frequency becomes larger to increase the feedback current to control the DC-DC converter. To pass the large feedback current, the photocoupler PC must have a large gain. The DC-DC converter employing the feedback control must have proper phase and gain (feedback gain) values, to stabilize the feedback loop. It is known that a gain of unity or larger and a phase of an integer multiple of 360 degrees cause an abnormal oscillation in a control system. Namely, excessively increasing the gain of the photocoupler PC increases a risk of abnormal oscillation.
As another related art, Japanese Unexamined Patent Application Publication No. 2005-39975 discloses a current resonant converter.